System and method for increasing bandwidth for digital predistortion in multi-channel wideband communication systems

ABSTRACT

A method of operating a communications system includes receiving a signal at a digital predistorter (DPD), introducing predistortion to the signal using the DPD, and converting the predistorted signal to an analog signal using a digital-to-analog converter having a first bandwidth. The method also includes amplifying the analog signal, sampling the amplified signal using an analog-to-digital converter having a second bandwidth less than the first bandwidth, and extracting coefficients of the DPD from the sampled signal.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/480,267, filed Sep. 8, 2014; now U.S. Pat. No. 9,197,259 B2, issuedon Nov. 24, 2015, which is a continuation of U.S. patent applicationSer. No. 13/625,760, filed Sep. 24, 2012; now U.S. Pat. No. 8,873,674B2, issued on Oct. 28, 2014, which claims priority to U.S. ProvisionalPatent Application No. 61/538,042, filed on Sep. 22, 2011. Thedisclosures of each of these applications are hereby incorporated byreference in their entirety for all purposes.

BACKGROUND OF THE INVENTION

The present invention generally relates to wideband communicationsystems using multiplexing modulation techniques. More specifically, thepresent invention relates to a method of increasing instantaneous oroperational bandwidth for digital predistortion linearization in orderto compensate for nonlinearities and/or memory effects in multi-channelwideband wireless transmitters.

The linearity and efficiency of radio frequency (RF) power amplifiers(PAs) have become critical design issues for non-constant envelopedigital modulation schemes with high peak-to-average power ratio (PAR)values. This has happened as a result of the increased importance ofspectral efficiency in wireless communication systems. RF PAs havenonlinearities which generate amplitude modulation—amplitude modulation(AM-AM) distortion and amplitude modulation—phase modulation (AM-PM)distortion at the output of the PA. These undesired effects may createspectral regrowth in the adjacent channels, as well as in-banddistortion which may degrade the error vector magnitude (EVM).Commercial wireless communication systems may for example employ abandwidth in the range of 20 MHz to 25 MHz. In such an example, thesespectral regrowth effects create an undesired impact over a frequencyband which is more than 100 MHz to 125 MHz wide. The potential impactsmay include inter-system and intra-system interference. Therefore, thereis a need in the art for improved methods and systems related tocommunications systems.

SUMMARY OF THE INVENTION

It is advisable to employ a linearization technique in RF PAapplications in order to eliminate or reduce spectral regrowth andin-band distortion effects. Various RF PA linearization techniques havebeen proposed in the literature such as feedback, feedforward andpredistortion. One of the most promising linearization techniques isbaseband digital predistortion (DPD), which takes advantage of recentadvances in digital signal processors. DPD can achieve better linearityand higher power efficiency with a reduced system complexity whencompared to the widely-used conventional feedforward linearizationtechnique. Moreover, a software implementation provides the digitalpredistorter with a re-configurability feature which is desirable formulti-standards environments. In addition, a PA using an efficiencyenhancement technique such as a Doherty power amplifier (DPA) is able toachieve higher efficiencies than traditional PA designs, at the expenseof linearity. Therefore, combining DPD with a DPA using an efficiencyenhancement technique has the potential for maximizing system linearityand overall efficiency.

Requirements for instantaneous bandwidth (for example, exceeding 25 MHz)for next generation wireless systems continue to increase, which meansthat the DPD processing speed needs to be increased accordingly. Thishigher processing speed may result in new digital platform designefforts, which can often take several months and significant staffresources and costs to complete. The higher processing speed can alsoresult in higher system costs and increased power consumption due tosampling rate increases for DPD in Field Programmable Gate Arrays(FPGAs), digital-to-analog converters (DACs), and analog-to-digitalconverters (ADCs). In addition, RF/IF filter requirements are morestringent, which will also likely increase system costs and complexity.Another typical result of having DPD with a wider instantaneousbandwidth may be increased memory effects. This may cause the DPDalgorithm to become much more complex and will take longer to design,optimize and test.

Embodiments of the present invention provide DPD linearization methodsand systems that provide wider bandwidth without adding a high degree ofcomplexity and cost.

Accordingly, embodiments of the present invention overcome thelimitations previously discussed. Embodiments of the present inventionprovide a method of increasing DPD linearization bandwidth withoutcostly modifications to an existing digital platform for multi-channelwideband wireless transmitters. To achieve the above objects, accordingto some embodiments of the present invention, a DPD feedback signal isemployed along with a narrow band-pass filter in the DPD feedback path.The embodiments described herein are able to extend the DPD bandwidthobtainable with an existing digital transmitter system, without changesin digital signal processing components, which could result in increasedpower consumption and/or cost.

Numerous benefits are achieved by way of the present invention overconventional techniques. As described more fully herein, the digitalfinite impulse response (FIR) filter characteristic for DPD output isimportant in order to avoid the overlapping of distortions, which cancause errors in an indirect learning algorithm based on DPD output andfeedback input. This can result in the need to utilize digital FIRfilters with a large number of taps. Embodiments of the presentinvention may provide for the removal of the digital FIR filter by usinga direct learning algorithm based on DPD input and feedback input.Accordingly, embodiments of the present invention may decrease thenumber of multipliers for multiband applications and even for singleband applications. Moreover, an analog filter characteristic in theanalog feedback path may also be provided in order to avoid theoverlapping of distortions, which can cause errors in the calculation ofcoefficients. Thus, embodiments of the present invention reduce oreliminate the need for one or more multi-pole ceramic filters, which arebig and expensive, replacing them with one or more less-stringentceramic filters that only removes aliasing in the feedback ADC.Accordingly, the digital FIR filter with a large number of taps can beinserted in order to avoid the overlapping. Furthermore, one filter canbe shared for multiband applications. Consequently, embodiments of thepresent invention can utilize a normal ceramic filter and one sharpdigital filter with a large number of taps. These and other embodimentsof the invention along with many of its advantages and features aredescribed in more detail in conjunction with the text below and attachedfigures.

BRIEF DESCRIPTION OF THE DRAWINGS

Further objects and advantages of the invention can be more fullyunderstood from the following detailed description taken in conjunctionwith the accompanying drawings in which:

FIG. 1 is a simplified flowchart illustrating a method of increasing DPDlinearization bandwidth according to an embodiment of the presentinvention.

FIG. 2 is a schematic block diagram illustrating a multi-carrierwideband power amplifier system according to an embodiment of thepresent invention.

FIG. 3 is a schematic block diagram illustrating a multi-carrierwideband power amplifier system according to another embodiment of thepresent invention.

FIGS. 4A-4D are graphs showing the DPD bandwidth characteristics for aconventional system.

FIGS. 5A-5D are graphs showing the DPD bandwidth characteristicsaccording to an embodiment of the present invention.

FIG. 6 is a plot showing spectral output response for a conventionalsystem employing DPD. and

FIGS. 7A-7C are plots showing spectral output response for systemsaccording to various embodiments of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

In general, the DPD techniques of the present invention can effectivelyimprove the adjacent channel power ratio (ACPR). However, DPDperformance suffers from the limited bandwidth associated with the speedlimitation of the ADC employed in the DPD feedback path. This ADC iscritical to processing the DPD feedback signals. Although modifying aproduct design to employ an ADC with a higher sampling rate would likelylead to enhanced DPD performance, that approach would increase thecomplexity and cost of the DPD function and would therefore result inhigher system cost. This is obviously an undesirable approach formeeting new and evolving system requirements. In order to overcome theselimitations, the present invention utilizes the bandpass characteristicof the duplexer associated with frequency division duplex wirelesssystems, so that the DPD is only required to provide distortionreduction over the reduced bandwidth of the PA output signals. Thesystem provided by the present invention is therefore referred to as anenhanced-bandwidth digital predistortion (EBWDPD) system hereafter.Embodiments of the EBWDPD system are illustrated with respect to theaccompanying drawings.

In conventional systems, the bandwidth associated with the DPD system istypically required to be five times the bandwidth of the input signal.For example, for a conventional system with a 20 MHz input signalbandwidth, the DPD function requires at least 100 MHz bandwidth for theDPD output and DPD feedback input, which means that feedback ADCsampling rate should be at least 200 Msps. This is a critical factor fora conventional DPD implementation.

FIG. 2 is a schematic block diagram illustrating a multi-carrierwideband power amplifier system according to an embodiment of thepresent invention. The system illustrated in FIG. 2 comprises digitalcomplex input samples 201 (with bandwidth of 20 MHz), digitalpredistortion circuits 202 (with bandwidth exceeding 100 MHz), a digitalfilter 203 with a similar bandwidth to that of the feedback band-passfilter 204 (FB BPF), digital-to-analog converters 205, an IQ modulatorshown as AQM 206, a power amplifier 207, a duplexer 208 (with bandwidthof 20 MHz), radio frequency down-conversion circuits 209 with a lowpower feedback RF band-pass filter 204 (RF FB BPF) for the outputcoupled at the output of the PA 207, and an analog-to-digital converter210 (with a bandwidth typically greater than the RF FB BPF bandwidthobtained by employing a sampling rate greater than two times the FB BPFbandwidth value) for the DPD feedback path. The RF FB BPF 204 filtersthe feedback signal to provide a signal characterized by a reducedbandwidth in comparison with the output of the power amplifier. DPDcoefficients are extracted from the feedback signal produced by the RFFB BPF 204, which has a reduced bandwidth associated with the filter204.

The DPD 202 introduces distortion components associated with the 3rdorder and 5th order expansion of the input signal, which causes the DPDoutput bandwidth to be larger than approximately 100 MHz based on a 20MHz input signal. In order to avoid instability of the DPD algorithm dueto inaccurate error calculation from the DPD output (with bandwidthexceeding 100 MHz) and feedback signal (with FB BPF bandwidth), the DPDoutput is filtered by a digital filter 203 having a bandwidth valuesimilar to that of the RF FB BPF 204. Embodiments of the presentinvention utilize an RF FB BPF 204 with a suitable bandwidth value asdescribed more fully in relation to FIGS. 7A-7C. The bandwidth of filter204 is less than the DPD bandwidth, which contrasts with conventionalsystems in which filter 204 would have a bandwidth equal to the DPDbandwidth. Additionally, the ADC 210 has a bandwidth associated with theFIR filter 203 in some embodiments, which is less than the DPDbandwidth.

It should be noted that in comparison with conventional systems, thebandwidth of various components in the multi-carrier wideband poweramplifier system illustrated in FIG. 2 are reduced, thereby reducingsystem complexity and cost. As an example, the digital filter 203 has abandwidth similar to that of the feedback band-pass filter 204 ratherthan exceeding 100 MHz based on the bandwidth of the digitalpredistortion circuits. The ADC 210 has a bandwidth typically greaterthan the RF FB BPF bandwidth obtained by employing a sampling rategreater than two times the FB BPF bandwidth value. Thus, embodiments ofthe present invention utilize components that operate at lowerbandwidths and sampling rates than conventional components in aconventional system, reducing the system cost and complexity.

FIG. 3 is a schematic block diagram illustrating a multi-carrierwideband power amplifier system according to another embodiment of thepresent invention. This embodiment shares some common features with thesystem illustrated in FIG. 2 as well as some differences. As illustratedin FIG. 3, the system includes a low power narrowband IF band-passfilter 301. Embodiments of the present invention provided by the systemillustrated in FIG. 3 may be easier and less costly to design andimplement using an IF BPF filter compared to using an RF band-passfilter. With an IF filter, the present invention is applicable tosystems employed with various applications based on the use of a commonIF frequency. As was the case for the embodiment shown in FIG. 2, thefeedback ADC following the IF FB BPF employs a sampling rate greaterthan two times the FB BPF bandwidth value for the DPD feedback path.This helps reduce the implementation cost while providing highperformance. The feedback loop provides inputs (e.g., a measure ofdistortion in the power amplifier 207) that are used to introducedistortion that compensates for the amplifier distortion.

The embodiments shown in FIGS. 2 and 3 may employ a digital filter 203characterized by a bandwidth that is less than the bandwidth (e.g., >100MHz) used in conventional systems. Additionally, the embodiments shownin FIGS. 2 and 3 may include either a low power feedback IF BPF or an RFBPF coupled to the PA output. Thus, filtering can be performed at RF orIF according to various embodiments of the present invention.

FIGS. 4A-4D are graphs showing the DPD bandwidth characteristics for aconventional system. The DPD bandwidth for conventional systems isrequired to be greater than 5 times the value of the input signalbandwidth. FIG. 4A shows the DPD input signal. FIG. 4B shows thefeedback signal, with distortion components (dark shading) over a fairlywide bandwidth of FB BW. FIG. 4C shows the DPD output signal withpredistortion components (based on the feedback signal) along with theFIR digital filter bandpass characteristic. The signal withpredistortion components has a bandwidth of slightly less than the DPDbandwidth. FIG. 4D shows the PA/duplexer output signal with distortionhaving been canceled. The data is included in the central spectral bandand distortion is illustrated in FIG. 4B and a distortion component witha 180 degrees phase shift (out of phase) is illustrated in FIG. 4C,resulting in cancellation of the distortion and the signal illustratedin FIG. 4D, with no significant out of band power. In some embodiments,the signal at the output of DPD 202 is similar to that illustrated inFIG. 4B.

As illustrated in FIG. 4D, the duplexer bandwidth is slightly greaterthan the bandwidth of the data spectrum. Embodiments of the presentinvention utilize the filtering properties of the duplexer 208 to assistin removing some of the out of band power from the spectrum. Because ofthe use of the duplexer, it is not necessary to correct across theentire bandwidth (e.g., FB BW), but only a portion of the bandwidth withthe duplexer providing a filtering function.

FIGS. 5A-5D are graphs showing the DPD bandwidth characteristicsaccording to embodiments of the present invention. As explained above inrelation to FIG. 2, the DPD bandwidth is associated with the FB BPFbandwidth, which is less than the bandwidth required by conventionalsystems. FIG. 5A shows the DPD input signal. FIG. 5B shows the bandwidthof the feedback signal after the FB BPF 204. As illustrated in FIG. 5B,the bandwidth of the feedback signal after the FB BPF 204 is reduced incomparison to the DPD bandwidth. Thus, referring to FIG. 2, RF feedbackband pass filter (RF FB BPF) 204 has a bandwidth as illustrated in FIG.5B. This bandwidth is reduced in comparison to the DPD bandwidth.

FIG. 5C shows the DPD output signal with predistortion components (basedon the feedback signal) along with the narrower FIR digital filterbandpass characteristic, compared to that for a conventional system. Thesignal with predistortion components has a bandwidth of much less thanthe DPD bandwidth. As illustrated in FIG. 5C, the predistortioncomponent 430 (see FIG. 4C) is greater than the predistortion component530. This results from the filtering properties provided by RF FB BPF204. It should be noted that the bandwidth associated with thepredistortion component 530 is much narrower than the DBD BW.

FIG. 5D shows the PA/duplexer output signal. In contrast with FIG. 4D,the Duplexer has a significant role in reducing output distortion welloutside the bandwidth of the input signal. Close to the respective bandedges of the desired signal, the DPD provides a substantial amount ofdistortion reduction. Thus, using the filtering properties of theduplexer enables compensation over a smaller range than otherwiseavailable. Close to the carrier, the out of band power (outside the dataspectrum) is substantially zero as a result of the digital predistortiontechniques used herein. Although some out of band power is present, themajority of the power is outside the bandwidth of the duplexer,resulting in the majority of the power being filtered by the duplexer.

FIG. 6 is a plot showing spectral output response for a conventionalsystem employing DPD. The results in FIG. 6 are for a conventional PAsystem without any FB BPF. The results are for a 4 carrier WCDMA inputsignal (with a total bandwidth of 20 MHz) and 60 W average output power.The bandwidth of the distortion is ˜100 MHz (i.e., 5 times the signalbandwidth). DPD reduces distortion more than 20 dB.

FIGS. 7A-7C are plots showing spectral output response for systemsaccording to various embodiments of the present invention. The spectrumshown in FIGS. 7A-7C illustrate DPD performance based on various valuesof FB BPF bandwidth (FIR filter 203) (25 MHz, 30 MHz and 40 MHzrespectively). With 25 MHz FB BPF bandwidth, the spectrum associatedwith DPD performance includes noise at a predetermined level. Systemsusing FB BPF bandwidths of 30 MHz and 40 MHz provide results for DPDperformance that are comparable to the DPD performance for conventionalsystems, while utilizing an ADC 210 having a much lower sampling ratethan the feedback ADC employed in a conventional system, which maybe >100 MHz. Additionally, embodiments of the present invention utilizea filter 203 that is characterized by much lower bandwidth than aconventional filter in a conventional system which has a typical valueof bandwidth greater than five times the signal bandwidth. The systembandwidth (i.e., 25 MHz) refers to the feedback loop and the bandwidthof RF FB BFP 204 in FIG. 2 or IF FB BPF 301 in FIG. 3.

Table 1 is a table showing Adjacent Channel Leakage Power Ratio (ACLR)performance for embodiments of the present invention, whose values aretaken from results of FIG. 6 and FIGS. 7A-7C. Table 1 is a table thatshows in various rows the ACLR performance of: PA system without DPD, PAwith conventional DPD approach, PA with DPD with 25 MHz FB BPF accordingto the present invention, PA with DPD with 30 MHz FB BPF according tothe present invention and PA with DPD with 40 MHz FB BPF according tothe present invention. Based on the data shown in Table 1, systemsutilizing a FB BPF with 30 MHz minimum bandwidth are able to achieveperformance similar to the conventional PA with DPD. Therefore, someembodiments of the present invention utilize a 30 MHz feedback pathbandwidth, meaning that a feedback ADC with a sampling rate of only 60Msps can be employed. This contrasts with conventional DPD systems thatrequire a feedback ADC with 200 Msps or greater sampling rate forsimilar performance.

In some embodiments, a 60 Msps feedback ADC is used for a 20 MHzinstantaneous input signal bandwidth and a Duplexer is used with 25 MHzbandwidth. In some embodiments, a Duplexer is used that has a bandwidthslightly larger than the instantaneous or operational input signalbandwidth. In some embodiments, the value of feedback bandwidth is setat a value approximately 20% greater than the instantaneous oroperational input signal bandwidth. In some embodiments, a system whichsupports a 60 MHz instantaneous or operational input signal bandwidthhas its value of feedback bandwidth set to 72 MHz, such as would resultfrom employing a feedback ADC with a 144 Msps sampling rate. Thus,embodiments of the present invention provide benefits (including reducedcost and complexity) not available using a conventional DPD systememploying a feedback ADC with a 250 Msps sampling rate, which is apopular choice for many conventional DPD systems.

TABLE 1 ACLR (dBc) @ ACLR (dBc) @ SYSTEM DESCRIPTION +5 MHz(+10 MHz) −5MHz(−10 MHz) PA without DPD −37.1(−38.8) −28.2(−30.37) Conventional DPD/−51.64(−51.83)/−52.29(−53.21) −50.38(−51.14)/−50.84(−52.57) System(25MHz) DPD(25 MHz)/System(25 MHz) −47.89(−45.6)/−48.72(−47.2) −46.8(−45.46)/−47.46(−47.01) DPD(30 MHz)/System(25 MHz) −50.85(−50.2)/−51.54(−51.75)  −50.0(−50.84)/−50.49(−52.23) DPD(40MHz)/System(25 MHz) −51.35(−51.45)/−51.99(−52.88)−50.33(−51.46)/−50.79(−52.85)

As illustrated in Table 1, the power amplifier without DPD has an ACLRvalue of −37.1 dBc and −28.2 dBc at +5 MHZ and −5 MHz, respectively.Using a conventional system, values of −51.64 dBc, etc. and −50.38 dBc,etc. are achieved. Utilizing embodiments of the present invention, asshown on the last three lines, values of −47.89 dBc, −50.85 dBc, and−51.35 dBc, respectively, are achieved. Thus, although performance isslightly degraded for the 25 MHz system of the present invention,performance improves for the 30 MHz system and is substantiallyequivalent for the 40 MHz system. Thus, embodiments of the presentinvention can utilize systems operating over a much narrower bandwidth(i.e., 40 MHz) than conventional DPD systems (i.e., 100 MHZ).

FIG. 1 is a simplified flowchart illustrating a method of increasing DPDlinearization bandwidth according to embodiments of the presentinvention. The method 100 includes receiving a complex input signal at aDPD (101) and introducing predistortion to the signal using the DPD(102). The method also includes filtering the predistorted signal usinga digital filter (103) and converting the filtered signal to an analogsignal (104). Filtering the predistorted signal can be performed over afilter bandwidth less than the bandwidth of the DPD, for example, over afilter bandwidth between 30 MHz and 50 MHz.

The method further includes quadrature modulating the analog signal(105), amplifying the modulated signal (106), coupling a portion of theamplified signal to provide a feedback signal (107), and filtering thefeedback signal using a band-pass filter (108). Filtering the feedbacksignal using the band-pass filter can be performed over a band-passbandwidth less than the bandwidth of the DPD, for example, the band-passbandwidth can be between 30 MHz and 50 MHz.

Additionally, the method includes downconverting the filtered feedbacksignal (109), converting the downconverted signal to a digital signal(110), and providing the digital signal to the DPD at its feedback input(111). Converting the downconverted signal can be performed at asampling rate less than twice the bandwidth of the DPD, for example, ata sampling rate is between 60 Msps and 100 Msps.

It should be appreciated that the specific steps illustrated in FIG. 1provide a particular method of increasing DPD linearization bandwidthaccording to some embodiments. Other sequences of steps may also beperformed according to alternative embodiments. For example, alternativeembodiments of the present invention may perform the steps outlinedabove in a different order. Moreover, the individual steps illustratedin FIG. 1 may include multiple sub-steps that may be performed invarious sequences as appropriate to the individual step. Furthermore,additional steps may be added or removed depending on the particularapplications. One of ordinary skill in the art would recognize manyvariations, modifications, and alternatives.

Although the present invention has been described with reference to thepreferred embodiments, it will be understood that the invention is notlimited to the details described thereof. Various substitutions andmodifications have been suggested in the foregoing description, andothers will occur to those of ordinary skill in the art. Therefore, allsuch substitutions and modifications are intended to be embraced withinthe scope of the invention as defined in the appended claims.

What is claimed is:
 1. A method of operating a communications system,the method comprising: receiving a signal at a digital predistorter(DPD) having a DPD bandwidth; introducing predistortion to the signalusing the DPD; converting the predistorted signal to an analog signalusing a digital-to-analog converter having a first bandwidth; modulatingthe analog signal using a modulator; amplifying the modulated signal;filtering the amplified signal using a duplexer having a duplexerbandwidth less than the first bandwidth; transmitting the filteredsignal using an antenna; sampling the amplified signal using ananalog-to-digital converter having a second bandwidth less than thefirst bandwidth; and extracting coefficients of the DPD from the sampledsignal.
 2. The method of claim 1, further comprising up-converting themodulated signal.
 3. The method of claim 1, further comprising filteringthe amplified signal using a bandpass filter.
 4. The method of claim 3,wherein the bandpass filter has a filter bandwidth less than the DPDbandwidth.
 5. The method of claim 1, further comprising downconvertingthe amplified signal.
 6. The method of claim 1, wherein converting thepredistorted signal to the analog signal is performed at a sampling rateless than twice a DPD bandwidth.
 7. A communications system comprising:a digital predistorter (DPD) configured to predistort an input signaland to produce a DPD output signal having a DPD bandwidth; adigital-to-analog converter having a first bandwidth, thedigital-to-analog converter configured to convert the DPD output signalto an analog signal; a modulator configured to modulate the analogsignal; a power amplifier configured to amplify the modulated analogsignal to produce a power amplifier output signal; a duplexer configuredto filter the power amplifier output signal to produce a filteredsignal, wherein the duplexer has a duplexer bandwidth less than thefirst bandwidth; an antenna configured to transmit the filtered signal;and an analog-to-digital converter having a second bandwidth less thanthe first bandwidth, the analog-to-digital converter configured toconvert the power amplifier output signal to a digital signal, whereinthe digital signal includes coefficients configured to be input to theDPD.
 8. The system of claim 7, further comprising a bandpass filterconfigured to filter the power amplifier output signal.
 9. The system ofclaim 8, wherein the bandpass filter has a filter bandwidth less thanthe DPD bandwidth.
 10. The system of claim 7, further comprising adown-converter configured to down-convert the power amplifier outputsignal.
 11. The system of claim 7, wherein the digital-to-analogconverter has a sampling rate less than twice the DPD bandwidth.
 12. Acommunications system comprising: a digital predistorter (DPD)configured to generate a DPD output signal based on an input signal, theDPD initially using a set of first DPD coefficients, wherein the DPD hasa DPD bandwidth; a digital-to-analog converter having a first bandwidth,the digital-to-analog converter configured to sample the DPD outputsignal to produce an analog signal; a modulator configured to modulatethe analog signal; a power amplifier configured to amplify the modulatedanalog signal to produce a power amplifier output signal; a duplexerconfigured to filter the power amplifier output signal to produce afiltered signal, wherein the duplexer has a duplexer bandwidth less thanthe first bandwidth; an antenna configured to transmit the filteredsignal; and feedback components having a second bandwidth less than thefirst bandwidth, the feedback components configured to sample the poweramplifier output signal to produce a digital signal, wherein the digitalsignal includes a next set of DPD coefficients.
 13. The system of claim12, further comprising an up-converter configured to up-convert themodulated signal.
 14. The system of claim 12, further comprising abandpass filter configured to filter the power amplifier output signal.15. The system of claim 14, wherein the bandpass filter has a filterbandwidth less than the DPD bandwidth.
 16. The system of claim 12,further comprising a down-converter configured to down-convert the poweramplifier output signal.
 17. The system of claim 12, wherein thedigital-to-analog converter has a sampling rate less than twice the DPDbandwidth.